1. Field of the Invention
The invention relates to a switching power supply device, which controls on/off switching of a switching element based on a voltage occurring in auxiliary windings provided in an isolation transformer, and which obtains a DC output insulated from a DC power supply.
2. Description of the Related Art
FIG. 13 shows technology of the prior art for this type of switching power supply device, with a circuit configuration which is substantially the same as that of the switching power supply devices disclosed in Japanese Patent Laid-open No. 2001-231258, FIGS. 1 and 10 (corresponding to U.S. Pat. No. 6,362,984) and in Japanese Patent Laid-open No. 2002-209381, FIG. 1 (corresponding to U.S. Pat. No. 6,483,722). In this circuit, by alternately turning on and off the switching element Q1 and switching element Q2 based on the voltage appearing in two auxiliary windings P2, P3 provided in the isolation transformer T1, a resonance voltage is caused to appear in the primary windings P1 of the isolation transformer. As a result, the resonance voltages appearing in two secondary windings S1, S2 provided on the secondary side of the isolation transformer are full-wave rectified by rectifying and smoothing circuits D1, D2, Co with a center-tap configuration to obtain a constant DC output. In this circuit in FIG. 13, a resonance voltage is caused to appear in the primary windings P1, and the series resonance operation of the leakage inductance and resonance capacitance Cr of the isolation transformer T1 is utilized.
In this type of switching power supply device, when the two switching elements Q1 and Q2 are turned on, by applying a gate voltage with the timing at which current flows in each of the body diodes, zero-voltage turn-on is possible, and turn-on losses do not occur. Further, voltages of elements Q1 and Q2 occurring at turn-off are clamped by the DC power supply voltage Vin, so that there is almost no surge voltage, and a low-loss, low-noise power supply device easily can be realized.
FIG. 15 shows other prior art for this type of switching power supply device, namely a circuit configuration substantially the same as the switching power supply device disclosed in FIGS. 2, 4 and 5 of Japanese Patent Laid-open No. 2004-153948, (corresponding to U.S. Pat. No. 6,917,528) described below. A difference between the circuits of FIG. 15 and FIG. 13 is that the resonance voltage occurring in one set of secondary windings S1 on the secondary side of the isolation transformer is half-wave rectified by the rectifying and smoothing circuit D1 and Co to obtain a constant DC output. In the circuit of FIG. 13, energy is supplied to the smoothing capacitor Co both during intervals in which Q1 is on and during intervals in which Q2 is on, but in the circuit of FIG. 15 energy is stored in the exciting inductance of the isolation transformer T1 during intervals in which Q1 is on, and during intervals in which Q2 is on the stored excitation energy is supplied to the smoothing capacitor Co. The circuit of FIG. 15 has the characteristic that the secondary-side rectifying and smoothing circuit can be simplified.
In the circuit of FIG. 13, the illustrated conventional switching power supply device having a control circuit 1, gate driving circuit 2 (such as a resistor), detection/adjustment circuit 4 and control circuit 5, the leakage inductance of the isolation transformer T1 is used as the inductance component causing resonance with the resonance capacitor Cr. However, the peak of the current flowing in the switching element Q1 must be suppressed, and because the leakage inductance value is of the order of several hundred microHenries, the separated winding construction shown in FIG. 14 is employed to adjust the inductance to the required leakage inductance (see FIG. 10 of Japanese Patent Laid-open No. 2001-231258 (corresponding to U.S. Pat. No. 6,362,984)).
In the case of a separated winding construction, in order to secure an adequate isolation distance (spatial distance, creepage distance) between the primary windings P1 and secondary windings S1 and S2, a separate barrier 12 must be provided at the bobbin 10. Because the resulting structure is complex, there is the problem that costs are comparatively high.
In the circuit of FIG. 15 on the other hand, during intervals in which the switching element Q1 is on, the peak value of the current flowing in the switching element Q1 is suppressed by the exciting inductance of the isolation transformer. As a result, the leakage inductance value need only be of the order of several microHenries to several tens of microHenries, so that a stacked winding construction such as is shown in FIG. 16 is sufficient. Because a stacked winding construction is simple, costs can be reduced as compared with a separated-winding isolation transformer.
Below, steady-state operation of the circuit of FIG. 15 is explained using the waveform diagram of FIG. 17. In FIG. 17, VQ1 and IQ1 are the voltage and current of switching element Q1, VQ2 and IQ2 are the voltage and current of switching element Q2, ID1 is the current of rectifying diode D1, and Im is the exciting current of isolation transformer T1.
Interval t4 to t1
In this interval, excitation energy is stored in the excitation inductance of the isolation transformer T1. IQ1 and Im have the same waveform. During the interval in which the body diode of the switching element Q1 is conducting and IQ1 is negative, when a gate voltage is applied to switching element Q1, switching element Q1 is turned on at zero voltage, and no turn-on loss occurs.
Interval t1 to t2
The switching element Q1 is turned off, and the excitation current is divided and flows to the parasitic capacitance (output capacitance), not shown, of switching elements Q1 and Q2. As a result, VQ1 rises and VQ2 falls, as shown in the figure.
Interval t2 to tr1
During the interval when VQ1 reaches the DC power supply voltage Vin and VQ2 reaches zero, the body diode of the switching element Q2 becomes conducting and IQ2 is negative. Moreover, when a gate voltage is applied to switching element Q2 the switching element Q2 is turned on at zero voltage, and no turn-on losses occur. The resonance current of the resonance capacitor Cr and the leakage inductance of the isolation transformer T1 flows in the switching element Q2. At the same time, a resonance current also flows in the rectifying diode D1, and energy is supplied to the smoothing capacitor Co.
Interval tr1 to t3
When the current of rectifying diode D1 goes to zero, the rectifying diode D1 undergoes reverse recovery and is turned off. The resonance current that had been flowing in the switching element Q2 also goes to zero, and IQ2 has the same waveform as Im. The rate of decrease of current when the rectifying diode D1 undergoes reverse recovery becomes more gentle with resonance operation, and the reverse recovery loss is slight.
Interval t3 to t4
When the switching element Q2 is turned off, the excitation current is divided and flows to the parasitic capacitance (output capacitance), not shown, of switching element Q1 and switching element Q2. As a result, VQ2 rises and VQ1 falls.
As explained above, in steady-state operation Q1 and Q2 are turned on at zero voltage during turn-on, so that low-loss, low-noise switching operation can be continued.
However, there exist switching power supply loads, which are switching loads that repeatedly alternate between light loads and heavy loads. An example of this is the backlight inverter or the like that drives cold cathode tubes used in the backlights of liquid crystal televisions and the like. Because cold cathode tubes cannot adjust the light emission intensity (brightness) through an applied voltage, the ratio of light emission intervals to extinction intervals is adjusted to control the brightness, and so a switched load results.
In the case of the circuit of FIG. 15, in order to suppress the drop in output voltage when the load increases rapidly, the resonance current must be increased rapidly; but at this time the switching element Q2 may turn off before the current flowing in the rectifying diode D1 reaches zero. Below, the waveform diagram of FIG. 18 is used to explain the problem that arises when an isolation transformer with the general winding construction shown in FIG. 16 is applied. In FIG. 16, the transformer has a winding construction in which primary windings P1, secondary windings S1, and auxiliary windings P2 and P3 are wound, in this order, on the bobbin 11.
Referring to FIG. 18, VGS2 is the gate voltage of switching element Q2. VP1, VP3 and VS1 are respectively the primary winding (P1) voltage, the voltage of the auxiliary windings (P3) that turn on and off the switching element Q1, and the secondary windings (S1) voltage of the isolation transformer T1. ID1 is the current in rectifying diode D1. VQ1, IQ1, and VGS1 are respectively the voltage, current, and gate voltage of the switching element Q1. Also, Vth is a threshold voltage for turning on and off the switching element Q1, based on the voltage appearing in the auxiliary windings P3.
In the case of the winding construction of FIG. 16, the auxiliary windings P3 are wound above the secondary windings S1. As a result, the degree of coupling with the primary windings P1 is weak, and the degree of coupling with the secondary windings S1 is strong. Hence, the voltage of the auxiliary windings P3 is substantially similar to the voltage of the secondary windings S1.
When the switching element Q2 turns off at time t3, VQ1 falls. The VQ1 voltage waveform falls due to resonance action between the excitation inductance of the isolation transformer and the combined capacitance of the output capacitances, not shown, of the switching elements Q1 and Q2. On the other hand, the voltage VP3 of the auxiliary windings P3 does not change until reverse recovery of the rectifying diode D1 at time tr2, and even when VQ1 begins to rise again due to resonance action, the switching element Q1 cannot be turned on.
When such action occurs, the switching element Q1 does not turn on at zero voltage. As a result, turn-on losses occur, and the efficiency of the switching power supply device is reduced. Alternatively, the switching operation may become unstable, and unpleasant sounds may be emitted from the isolation transformer.
Next, a separate problem is explained. When using a power supply startup method in which a resistance connected to the DC power supply causes the gate voltage of the switching element Q1 to be raised, providing an opportunity for the switching element Q1 to turn on, the auxiliary windings voltage undergoes high-frequency oscillation near the threshold of the switching element Q1. As a result, there are cases in which an adequate output voltage is not generated and startup fails. This is illustrated in FIGS. 2, 4, and 5 of Japanese Patent Laid-open No. 2004-153948 (corresponding to U.S. Pat. No. 6,917,528),
Hence an object of this invention is to provide an inexpensive switching power supply device which, even in a mode in which the switching element Q2 turns off before the current flowing in the rectifying diode becomes zero, turns on at zero voltage and prevents a decline in the efficiency of the switching power supply device Such a device also should stably continue switching operation and eliminate unpleasant sounds from the isolation transformer. Startup failures should not occur.